The electromagnetic actuation of the valves requires electrical energy. This electrical energy is taken from the network on board the vehicle. The performance of the electromagnetic valve system involves minimizing the electrical energy consumption on the on-board network. In practice, the power available at the output of the crankshaft is equal to the total power developed by the heat engine minus the power needed for the correct operation of its auxiliaries (actuation of the valves, driving of the water and oil pumps, etc).
Generally, a positive- or negative-current-controlled electrical machine is often controlled by H-configuration power bridges, also called “four quadrant” bridges (or “full-bridge chopper”) that are single-phase or polyphase depending on the structure of the machine. The term “four quadrant” bridge should be understood to mean a bridge control that works on all the quadrants of the voltage-current characteristic. Generally, the control electronics slave the current to a set point by applying, to the power bridges, a PWM (pulse width modulation) type control with fixed frequency. An example of the electronic structure of a “four-quadrant” bridge 1 is represented in FIG. 1. The bridge 1 comprises:                four terminals 5, 6, 7 and 8,        a DC voltage source 9 (this may be, for example, a battery, a DC-DC power converter or an AC-DC power converter) connected to the first and second terminals 5 and 6 of the bridge 1,        an electrical control member 10 (such as an electrical load with variable inductance) used to control an actuator and connected between the third and fourth terminals 7 and 8 of the bridge circuit 1,        a first switch C1 connected between the first and third terminals 5 and 7,        a second switch C2 connected between the first and fourth terminals 5 and 8,        a third switch C3 connected between the second and third terminals 6 and 7,        a fourth switch C4 connected between the second and fourth terminals 6 and 8.        
The bridge 1 therefore comprises two legs B1 and B2 respectively formed by the switches C1 and C3 in series and by the switches C2 and C4 in series.
The power switches C1 and C4 may be MOSFET (metal oxide semiconductor field effect transistor) transistors or IGBT (insulated gate bipolar transistor) transistors.
The technique most commonly used for its robustness is controlling the average current by using a control of the switches by pulse width modulation (PWM). Its greatest advantage is its immunity to noise. In practice, the regulation loop does not need an instantaneous current but a filtered value. This filtering introduces a delay which can prove highly restrictive when a high bandwidth is sought. In the case of a “camless” system actuator, the value supervisor calculates a voltage set point V* which must be applied to the terminals of the magnetic circuit. The pulse width modulation strategy translates the voltage set point into an instruction to close/open the electronic switches of the power electronics (in this case four of them). The power electronics apply the commands of the PWM strategy while observing its own constraints (management of dead times, whole times, etc). If Sci is used to denote the switching function of the switch Ci, we obtain:
      Sc    i    =      {                                        0            ⁢                                                  ⁢            if            ⁢                                                  ⁢                          C              i                        ⁢                                                  ⁢            is            ⁢                                                  ⁢            open                                                            1            ⁢                                                  ⁢            if            ⁢                                                  ⁢                          C              i                        ⁢                                                  ⁢            is            ⁢                                                  ⁢            closed                              The controls for two switches of one and the same bridge leg (Ci and Ci+2) are antagonistic to avoid causing a power supply short circuit. It is thus sufficient to give the trend over time of two switching functions (one for each leg) to determine the voltage Uact at the terminals of the magnetic circuit 10 according to the DC voltage UDC. For convenience, the switching functions of the high switches are routinely chosen. By neglecting the voltage drop in the conducting elements and the switch opening and closing times, the following can be deduced:Uact=(Sc1−Sc2)·UDC In the interests of simplicity, the time trends of the switching functions Sc1 and Sc2 are determined by comparing the value of the standardized voltage set point v* to a single triangular carrier Vp(t) of frequency fPWM. The triangular function may be any function that takes values between a minimum value Vpmin and a maximum value Vpmax. The value of the standardized set point voltage v* is then:
      v    *    =                                          V                          p              ⁢                                                          ⁢              max                                -                      V                          p              ⁢                                                          ⁢              min                                                2          ·                      U            DC                              ·              (                              V            *                    -                      U            DC                          )              +          V              p        ⁢                                  ⁢        max            By defining the function sign(x) by:
      sign    ⁡          (      x      )        =      {                                                      0              ⁢                                                          ⁢              if              ⁢                                                          ⁢              x                        ≤            0                                                                          1              ⁢                                                          ⁢              if              ⁢                                                          ⁢              x                        >            0                              the switching function Sc1(t) is determined as being:Sc1(t)=sign(v*−Vp(t))The function Sc2(t) is the complementary function of Sc1(t) that can also be calculated by the following formula:Sc2(t)=sign(Vp(t)−v*)
FIG. 2 graphically represents the determination of the switching functions Sc1(t) and Sc2(t). Thus, the first curve represents the trend as a function of time of the standardized set point voltage v* and of the triangular carrier Vp(t). From this, the second curve is deduced, representing the trend of Sc1 as a function of time and the third curve, complementing the second, representing Sc2 as a function of time. The fourth curve represents the trend of the voltage Uact across the terminals of the magnetic circuit 10 which varies from +UDC to −UDC and has an average value V* over a chopping period fPWM. Since the voltage Uact can take only two distinct values, this is referred to as a two-state PWM strategy.
FIG. 3 illustrates the switchovers induced by this type of PWM strategy on a four-quadrant bridge 11 comprising four switches AH, BH, AL and BL respectively identical to the transistors C2, C1, C4 and C3 of the bridge 1 as represented in FIG. 1 for the control of a load 10.
The bridge 11 has two possible states:                a magnetization state (with a voltage VL=+Udc applied to the actuator) in which the switches BH and AL are closed and the switches AH and BL are open,        a demagnetization state (with a voltage VL=−Udc applied to the actuator) in which the switches BH and AL are open and the switches AH and BL are closed.        
Implementing such PWM control for a four-quadrant bridge does, however, raise certain difficulties.
A first difficulty relates to the switching losses. When a switch changes state (transition from open to closed or transition from closed to open), it originates losses due to the simultaneous presence of current passing through it and a voltage at its terminals. The energy then dissipated depends on the value of the chopped current Iact, of the DC voltage UDC and of the speed of switching (the switching times are, for example, set by the value of the gate resistance of the MOSFET transistors used). Thus, for each chopping period, there are two openings and two closures regardless of the direction of the current on one of the two switches of each of the bridge lengths. This double switching for each chopping period obviously causes losses that become all the greater as the frequency increases. It will be noted that it is important to reconcile efficiency and bandwidth. In the case of PWM control, the PWM frequency is generally some tens or even hundreds of kilohertz. At these high frequencies, the switching losses predominate over the other conduction losses. It will be noted that, in the transition from a magnetization state to a demagnetization state, of the four switching operations for each chopping period, two are hard switching operations and two are soft switching operations: in other words, the first step is to open the two transistors that are initially closed (hard switchovers) then, after a dead time, the two initially open transistors are closed (soft switchovers). Having two switches of one and the same leg closed simultaneously is thus avoided. During the dead time (before the soft switchovers), the diodes (called “freewheeling”) intrinsic to the MOSFET switches conduct and thus make it possible to keep the MOSFET potential close to zero during the soft switchover.
Moreover, the repeated switchovers do not only affect the efficiency of the control electronics, but also the efficiency of the electrical machine that they drive. The chopping of the voltage generates high-frequency harmonics which induce losses in the electrical machines. These electrical machines generally consist of magnetic materials that favor eddy currents (iron-Si for example). The induction generates an induced voltage in the plates which, depending on their resistivity, creates often significant eddy currents. Although the plates are finely cut and insulated from one another, the currents that flow therein generate losses through the Joule effect. As mentioned above, the fourth curve of FIG. 2 represents the voltage Uact across the terminals of the load 10 as represented in FIG. 1 with a voltage source 9 having a value UDC. The control of the bridge is of PWM type with a duty cycle α. The voltage UDC, disregarding the voltage drops due to the resistances of the switches, is applied to the load 10. Whatever the duty cycle, if the latter is constant, the effective value (Ueddy) of the voltage applied to the load 10 is equal to the voltage UDC (which represents the peak voltage of the voltage Uact): Ueddy=UDC. Finally, the losses generated by the eddy current whose frequencies are above the chopping frequency and for which a constant duty cycle can be considered, are substantially proportional to the peak voltage: Peddy=k·Ueddy=k·UDC.
Furthermore, the control of electronics, and more particularly their switches, generate high-frequency common mode currents because of the chopping. The load generally has a capacitive coupling relative to earth. For example, in the case of a wound electrical machine, there is a significant coupling between the winding subjected to the common mode voltage and the frame linked to earth. Common mode currents are thus generated and they are looped by the power supply. These high-frequency current loops are responsible for the electromagnetic radiation that is likely to have an impact in terms of compliance with current EMC (electromagnetic compatibility) standards.